I have seen a number of low output PCB etched spiral Tesla coils on YouTube and wondered if the they could be driven to produce the larger outputs of traditional cylindrical coils. After designing my McTesla Tesla coil, I decided to try driving a PCB etched coil with my non resonant half bridge driver circuit. I first etched my own home brew coil which was 150 turns with an 8 mil trace spaced 8 mils. This worked with modest output but the frequency was higher than I desired at more than 2 MHz. So I broke down and had a commercially etched board made which was 6 inches square and has about 240 turns with a 6mil trace spaced 6 mils.
This coil performs well and depending on your top load resonates at anywhere from 900 kHz to 1.2 Mhz. The driver circuit is essentially the same as in my other designs. There are minor tweaks of the phase shifting network in the feedback path because of the higher frequency but everything else is the same. My top load is a 3D printed toroid which was wrapped in aluminum tape. The primary is one turn of etched on the bottom layer. Though the coil works quite well, it has to be carefully structured to avoid inter coil arching and breakdown.
My other coil designs utilizing this half bridge driver can achieve 9-10″ streamers. Surprisingly, the PCB etched coil can produce 12 inch streamers. To achieve such high output without the coil breaking down; I had to encapsulate the secondary in two part epoxy compound. Also the center post, connected to terminal, is screwed down to the secondary center with a nylon screw. A brass washer soldered to the center is the secondary output connection. There can be no metal screw passing through the PCB or it will fail to the primary or arc across the secondary on the top. Two part electrical potting compound works well to protect the coil and is easy to apply. It will harden in about an hour in a warm oven (170F). Both the top and bottom are potted. This coil uses the same interrupter circuit as the McTesla. The break out point is critical and the mosfets will immediately explode if the breakout point is not present. The breakout point can point up or to the side, but you have to be careful to make it extend an inch or more out from the terminal to provide adequate break out.
The McTesla is small 1000 watt half bridge Tesla coil with 3D printed forms for the terminal, secondary and the enclosure. The plastic used was PETG, which has low dielectric loss. The primary is 4 turns of 16 gauge automotive hookup wire(thick jacket) separated from the secondary about 1/8″ by means of 3D printed spacers. The power is provided directly from the mains through a half wave rectifier. The actual foot print is for a bridge rectifier but with a jumper wire you can fit the selected single diode on the board. Either will work fine. I get a little more crackle out of the half wave version so I just left it in place.
The secondary is 5″ tall and 3′ in diameter at the bottom and 5″ in diameter at the top. This alters the distribution of inductance of the coil and all but requires you have a substantial top load to suppress corona on the upper part of the winding. I get larger arcs with the conical secondary than an equivalent cylindrical secondary…maybe an inch or so more spark. I need to analyze why this is the case in more detail. The math shows a drum coil with equal diameter and height produces optimal Q. However, I think I am getting better primary coupling and impedance match allowing more energy couple to the secondary. It is wound with 700 turns of 33 gauge wire and resonates at 350 kHz. I tried 27 gauge wire and it worked fine. It resonated at 600kHz and streamer length was a couple inches less. Anything between 33-28 gauge will work well.
I have tried all sorts of different finishes for the secondary, including: polyurethane, and two part epoxy, ultimately, I ended up using 3 coats of spray shellac. It dries very quickly and you can apply 3 coats within a couple of hours. I found that when the coil is properly tuned and primary coupling is correct, a heavy insulative finish is not required.
The half bridge architecture I used for this is described in more detail in a previous blog post: https://wordpress.com/post/circuitsalad.com/1875 . The main feature is the inclusion of a simple passive adjustable phase shift network in the feedback path. This allow one to minimize switching losses by minimizing current and voltage overlap in the mosfets. The end result is the FETS are less stressed and need less heat sinking. In this design, I have made some minor changes from the previous circuit. I am now using the IXTQ52N30P 300V@52 amp mosfet instead of the IRFP4229. The first IRFP4229’s I got worked great but then the next two I ordered failed quickly. As well, they seem scarce and hard to source so I thought it best to try another. The IXQ52N30P is near indestructible and I have yet to blow one. I simplified the modulation input circuit and decided to remove the diodes I was using across the gate drives. The diodes were in place to ensure the mosfets were not both conducting at the same time but I found when properly tuned, this not needed. I also used terminal blocks for the mosfet connections for easy experimental swapping or replacement . You can see this on the picture of the circuit board below. The driver board uses a 200:1 current sensing transformer for magnetic feedback to generate self excitation. This feedback signal is then phase shifted with a two stages of RC low pass filtering. One stage is fixed and the other adjustable. The adjustment range is from about 1 MHz to 200 kHz and is very forgiving. Optimal adjustment can be achieved by use of an oscilloscope, but simply tuning for the largest spark and then reducing the resistance of the POT slightly works well too.
It includes native expresspcb cad files, schematics, gerbers, 3D print files and software development files
If anyone is interested in building this device everything you need is provided in the download. I am sure there will be questions which I am happy to answer. The interrupter is not required and can be replaced with a momentary switch or any other modulated switch closure. Some of the component choices are non-critical; such as 1N5819 diodes. The UCC27322 can be used instead of the UCC27321. The gate drive transformer can be any small 73 type ferrite material balun core or toroid. Other current sensing transformers will work but the CS1200L is a good choice and available. Certain IGBT’s will work well as well as other mosfets. You may need more heat sinking depending on the device selected. The AC to DC 12V module is a cheapo pcb module from Amazon. https://www.amazon.com/dp/B07SJRX9R6?psc=1&ref=ppx_yo2_dt_b_product_details
The Interrupter circuit is based on a PIC microcontroller and uses a simple opto- isolator as a transistor switching output. The OLED display used is a common 128×64 SSD1306 type. Ironically, I didn’t use the PWM output on the chip and created my own in software….but you easily could. I have it setup to provide 4 different duty cycles up to 50% , which can be adjusted by means of the two buttons. A rotary encoder sets frequency from 1hz -220Hz. I have driven the coil with 20Khz and it works. The momentary switch on the encoder is used to turn on and off the device. The software was written in C with the MikroC pro compiler. Next, I will make a PWM modulator to play music!
I decided to try using IGBT’s in my optimized half bridge Tesla Coil driver circuit. The results were good but the IGBT’s did have more heating than the very low RDS on mosfets such as the IRFP4229. I was able to phase shift the drive feedback such that zero crossing switching was achieved, but the Collector to Emitter voltage of approximately 1 volt @ 10 amp current produces 10 watts of dissipation, whereas the IRFP4229 RDS of. 04 ohm yields only 4 watts of dissipation @ 10 amps. So for a low power coil the FET is a better performer. At much larger currents, the IR dissipation the Mosfet grows exponentially, so in DRSSTC designs the IGBT would be a better choice as its dissipation grows linearly. My test coil operates at nearly 400KHz. This is a very high frequency for IGBT’s. The main issue with IGBTs is that they are slow to turn off when driven to saturation(switching). I chose the STGW35HF60W 600V, 35 amp ultra fast IGBT for my experiments. It is optimized to minimize turn off delay and residual tail current. It worked well at this frequency but the price you pay is a higher Collector to Emitter saturation voltage than you get with slower IGBTs. So again this is a trade off. Over all these IGBTs work well in the circuit and while they do get warmer than the Mosfets, they do not overheat with the small heat sinks and switching losses are low. They seem tolerant of abuse and I have not blown any up yet!
Lately, I have been experimenting with solid state Tesla Coil design ideas and after many destroyed mosfets, I have developed a reliable and efficient conventional solid state half bridge Tesla Coil driver. The topology is a basic half bridge so the primary is not resonant. This choice was made so I could operate it in continuous mode when desired. It is designed to operate with 120 VAC. The circuit is self exciting, utilizing current feedback from the secondary by means of a small 200:1 coilcraft current sensing transformer. The driver is efficient requiring minimal heat sinking of the mosfet switches. As such, the driver can be run for long intervals. The limitation on run time is constrained by the coils(secondary and primary ) and the electrolytic power supply capacitors heating up. I can generate 11 inch streamers on my 5″ diameter x 7″ tall secondary with a 6″ toroid terminal. I can just change out the different secondary and primary combinations I want to test, as the driver will easily operate from 100Khz to over 1 MHz. Depending on the frequency, you may need to change one resistor. The driver board is only 3.6″x 3.6″ and including simple stick on heatsinks for the mosfets.
Features that improve efficiency and reduce mosfet heating:
Properly designed gate drive: I used an appropriately sized and correct ferrite material xfmr core for the drive transformer. The core I used is very small (.3 inches) and was cannibalized from a small pulse xformer obtained from Mouser(photo below). There were no cheap small cores available on Mouser so i just found a small pulse isolation transformer that was inexpensive and I could get the core from. If you prefer, you can purchase stand alone core. Type 73 material is a good choice and a core no bigger than a .5 inch will be fine for this circuit. I cannibalized the core out of the housing and re-wound it with a trifilar 28 gauge winding of 5 turns. I also placed schottky diodes across the 10 ohm gate resistor providing for slightly faster off time than on time, This ensures minimal shoot through(both mosfets on at the same time). The drive is clean with no ringing and fast rise and fall times(pictures below). This reduces switching losses.
Choice of mosfet: I wanted to select a low cost and high performance mosfet for the circuit. There are two I found that work well. The basic criteria are : > 250V breakdown rating @ > 20 amp current capacity. Low output capacitance, gate capacitance and on resistance were also important. The IRFP4229 is the ideal candidate. It has a breakdown of 250V. It can handle 40 amps of current, and has total switch delay of of 100ns(very fast). The on resistance is .04 ohm; so IR losses are low. A similar device, only slightly slower and with a slightly greater on resistance is the STW46NF30. Both were evaluated and worked well. The cost of both of these mosfets is around $3 each.
Adjustable phase shift control in feedback loop: A substantial improvement in output and efficiency was obtained by placing a simple low pass RC phase shift network in series with the secondary feedback path. It was observed that when my test coil was driven from an external source, the input drive was approximately 90 deg out of phase with the secondary output. This was the case when the coil was tuned for maximum output. This amount of phase shift will not support oscillation properly. Forcing the coil to self excite will not produce optimal output unless some sort of phase shifting is employed. The notion that a self excited tesla coil will inherently tune itself to best performance is incorrect. Adjusting the input /output the phase relationship maximizes coil output but also lets one adjust the drain voltage/current relationship such that zero crossing switching can be obtained. This is when the current and voltage across the switches do not overlap. This greatly reduces power losses across the switch. The phase shift network consists of two simple RC low pass filters in series. One with a fixed resistor and the other is a variable resistor to allow adjustment. The low Q nature of the RC filter provides optimal tuning over a wide range and is easy to adjust and is stable. The values for the filter as shown in the schematic are easy to calculate. Simply pick a resistor value(10k for example), then use the approximate value of operating frequency(+-20% is fine) to determine the capacitors required. The value of capacitance is correct when the reactance of the capacitor at the given frequency is the same as the resistor value(10k). Or chose a capacitor value such as 47pf. With this value, the reactance at 650KHz is 1/2Pi(650KHz)(47pF), which is 5212 ohms. So you could use a 4.7k resistor for one RC stage and a 10k to 50k pot for the second stage. In my example coil, the frequency is around 400KHz and so I used 10k with 47pF and a 25K Pot with a 47 pF for the second stage. The low pass characteristic of the phase shifting also has the benefit of reducing higher frequency noise and parasitics in the feedback loop. The result of this simple addition to the circuit is substantial and when you tune the variable resistor you can see the output significantly increase and decrease. You can monitor the drain voltage and current to adjust phase for optimal performance. Alternatively, I have found that you can simply adjust for longest streamer length and then add just a tiny bit more phase shift to optimize switching efficiency.
I have really enjoyed my FV-1 based mini SDR radio but it has one problem…its too small! I made the thing so small; it’s hard to operate and assemble. So I decided to make it a little larger, allowing for all the controls to be larger and more spaced out. It has a larger display and the circuit board layout provides for the switches, encoder, volume control and display all to be soldered directly to the board. All of the circuit components are now on the top side of the board as well. Along with these physical changes, I made some minor circuit changes. These include: some component value changes, a different microcontroller, clocking the FV-1 at 48Khz with the third output of the SI5351 and adding a on/off power circuit which utilizes a momentary switch instead of a latching one. It now has a built in flat pack lithium ion battery that can be USB charged. I also refined the DSP demodulators and I am now utilizing a weaver demodulator for USB/LSB.
I started using a simple graphically based CAD tool(SPINCAD) to develop the demodulator DSP code for the FV-1, and I have been able to improve my demodulators algorithms. The CAD tool is free and is Java based. It runs natively in windows with Java installed. You can wire together functional building blocks and generate the required hex code for the FV-1 without writing any assembly code.
This is an example of a high efficiency QRP transmitter designed to work at very low supply voltages (3v-5v). It can produce 2 watts a 4 volt supply @ 70% efficiency. It uses small, inexpensive switching mosfets. The primary requirement for these mosfets is low output capacitance, a VDS of >20V, a logic level VGS and a drain current rating of a couple amps. There are many devices that will work. Unlike a Class E amplifier, this design requires no special alignment, providing for multiband operation easily. Only the output filter consisting of a L Network and Pi network in series need to be changed for a given band. It is tolerant of all kind of load conditions including infinite Z and maintains efficiency when poorly matched. While this circuit utilizes a microcontroller, display and clk generator, the logic buffer can operate from any oscillator source so the amplifier can be adapted to simpler designs.
Below is a video showing the restoration of a 1966 Fender Deluxe Reverb Amp that has sat unused since the 70’s and was barely used over its life. Not much had to be done so almost everything is original…including the tubes! It sounds fantastic and looks like new.
I decided to try using a small wide band E field type antenna with my newest receiver design…the Mini SDR and the results have been gratifying. There many useful articles describing this type of antenna; so I won’t go into much detail about how it works. More or less it functions as a capacitive E field probe and therefore is very sensitive to EMI. However, if placed outside away from house wiring and provided with a modest local ground reference..the antenna is a good performer. The classic circuit uses a JFET source follower and a BJT follower stage to provide impedance transformation of the Hi Z capacitive terminal to a 50 ohm Z drive for transmission line. This circuit works fine but has some drawback, namely requiring 65mA of current and having a somewhat large input capacitance, which reduces performance with frequency. I decided to use a wide bandwidth op amp to simplify the circuit, reduce current, and provide a little voltage gain. The op amp I chose was one I have used before for RF amplification..the LT1818. When choosing an op amp for such an application..there are a few important criteria to focus on:
1. current noise: Unlike low impedance topologies where voltage noise and resistor thermal noise will dominate, having a Hi Z input will make the current noise the dominant source of amplifier noise.
2. Bandwidth: you need a wide bandwidth on the order of hundreds of MHz or more to provide the required frequency response up to 30Mhz. This is especially true for voltage feedback op amps, where the phase shift compensation rapidly reduces performance over frequency.
3. Slew rate: You want the largest slew rate you can get to reduce distortion and IMD products.
4. Input bias current/ input Z/ input capacitance: You need low input capacitance so as to not to create a lossy divider with the antenna terminal. You want low input bias current and high input Z to not load down the terminal. If the input bias current is too high , then you need a low value bias resistor which loads the terminal.
5. Low output impedance: To drive 50 ohm Z and minimize distortion.
The LT1818 has excellent specs with regard to all of these criteria. It can operate on 3v-12v and requires only 9mA of current to operate. The Amplifier will operate from VLF to beyond 30Mhz with no change in performance.
The antenna is powered via a power splitter connected between the receiver and antenna. This is the purpose of L1 to isolate the DC power from the RF output from the antenna.
I 3D printed antenna capsule from HIPS, which is a low RF loss material and used a 3″ square of PC board to create the capacitive antenna terminal. The printing was done at 50% density so it’s a very light, low dielectric loss enclosure.
Go here for the most up to date circuit /firmware mods: Design Updates
This is a revised version of my FV-1 based SDR. I replaced the CS2100 clk generator with the Si5351 clk generator. The Si5351 has some advantages over the CS2100, namely you can generate quadrature clks directly. This simplifies the hardware design and improves the quadrature accuracy. The sideband rejection in LSB/USB modes is impressive..somewhere around 60 db as best I can measure. The DSP processing is accomplished by the use of a FV-1 audio processor. The device makes the base band signal processing a snap. It requires some code to be loaded on a EEprom but the circuitry is simple and allows for up to 8 selectable programs. I created three: AM/USB/LSB . The FV-1 provides for three analog POT inputs to control any parameters you choose. Gain, variable filter bandwidth and depth, AGC are some examples of adjustable parameters if you desire. I kept it simple and created fixed band pass filters to taste. I did use one of the controls for AF gain. The design has no tuned circuits or band pass filters but they could easily be added. It works just fine without them. Occasionally, I come across a ghost signal from harmonic mixing, when tuning, but not enough to matter. The design uses an OLED display and a rotary encoder for tuning. The frequency coverage is from 2.7 Mhz to 25Mhz. The bottom limit is created by the inability of the Si5351 to support quadrature below this frequency. Although I have improved my DSP programs for the FV-1 and have developed new display drivers and the new code for the Si5351, useful detail about using the Fv-1 can be found in my original design from a few years ago: https://circuitsalad.com/2015/06/19/comming-soon-stand-alone-software-defined-radio-baseband-demodulator-no-computer-required/
Schematic: Updated 05/17/2020
The design uses a LT1818 or THS4304 low noise op amp as an RF input with gain. It provides a constant and reliable resistive Rf termination for the sampling detector. This allows for random antennas to be used without adversely affecting the input termination to the detector. All the code to operate the main processor(display/clk generator/tuning, band select and receive mode) was written in MikroC which is a C compiler for PIC and AVR processors. The generation of quadrature signals out of the Si5351 is not difficult to implement once you know how but..figuring that out took me a couple weeks of experimentation! You can connect switches, the encoder, volume pot and display directly to the main board for operation but I created a secondary board to mount the display and encoders. Instead of an analog pot and selection momentary switches, I used another microcontroller and two encoders(with one built in momentary push switch each) to create all of the switching signals, gain control, etc. This allowed me to have just two controls for all features. The controls include: tuning, audio gain, mode, and tuning step. Tuning resolution is from 1Hz to 100KHz . For fun, I made the output of the FV-1 differential into the audio amp. This is not necessary.
Here is a link to all the files used to build this radio in a zip file(updated 2/07/20):
The schematic and PCB was done with express pcb freeware. The C compiler used was MikroC, and FV-1 assemble was built in SpinAsm which is free and available from Spin Semiconductor(who makes the Fv-1). The gerber files provided were created for OSH park. I had my boards etched by them. If anyone is interested in building this radio or leveraging elements of the design. I can answer questions.
Misc Notes: I use a 16650 3.7v lithium rechargeable battery to power the radio. The current draw is about 100 mA with audio. The radio works even when the regulators drop out so it will work at 3 v.
The enclosure is a machined aluminum 1590A style hammond box which you can buy on Ebay from alpinetech. They are $14.00 which is pricey but they are not cast. The quality is much nicer and you can anodize them. It’s a different topic but home anodizing of aluminum is easy…and I do it with all my enclosures now. In this example, I anodized twice to create the base blue color and then the labeling as well. It looks really clean with this method. The nice thing about anodizing is if you make a mistake, it’s really easy to go back and redo the process.
Designers will note that the resistive terminations on the input RF OP amp contributes to the noise figure of the radio. As a practical matter. a negative impact on performance is not noticeable because of atmospheric noise in the shortwave bands. For the best performance…no front end circuitry or a different front end input amplifier should be considered. Note that the op amp serves to bias the analog switches to half supply; so this bias must be provided to the sampling detector if the input termination is modified. R10 set the impedance of the sampling detector, conversion gain, and low pass roll off. The schematic shows a value 0f 210 ohms…I think I am using 100 ohms actually now…which works well.
If you want quadrature out of the Si5351 below 3MHz you can create two outputs with 0 deg offset with one output at F and the other at 2F. You can then drive an analog mux with those signals and generate quadrature sampling for low frequency applications. Just note the output sequence of the samples change so you have to flip two outputs of the detector.